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LT3474/LT3474-1 Step-Down 1A LED Driver FEATURES
n n n n n n n n n n n n
DESCRIPTION
The LT(R)3474/LT3474-1 are fixed frequency step-down DC/DC converters designed to operate as constant-current sources. An internal sense resistor monitors the output current allowing accurate current regulation, ideal for driving high current LEDs. High output current accuracy is maintained over a wide current range, from 35mA to 1A, allowing a wide dimming range. Unique PWM circuitry allows a dimming range of 400:1, avoiding the color shift normally associated with LED current dimming. The high switching frequency offers several advantages, permitting the use of small inductors and ceramic capacitors. Small inductors combined with the 16-lead TSSOP surface mount package save space and cost versus alternative solutions. The constant switching frequency combined with low-impedance ceramic capacitors result in low, predictable output ripple. With their wide input range of 4V to 36V, the LT3474/ LT3474-1 regulate a broad array of power sources, from 5V logic rails to unregulated wall transformers, lead acid batteries and distributed power supplies. A current mode PWM architecture provides fast transient response and cycle-by-cycle current limiting. Frequency foldback and thermal shutdown provide additional protection.
True Color PWMTM Delivers Constant Color with 400:1 Dimming Range Wide Input Range: 4V to 36V Up to 1A LED Current Adjustable 200kHz-2MHz Switching Frequency Adjustable Control of LED Current Integrated Boost Diode High Output Current Accuracy is Maintained Over a Wide Range from 35mA to 1A Open LED (LT3474) and Short-Circuit Protection High Side Sense Allows Grounded Cathode Connection Uses Small Inductors and Ceramic Capacitors LT3474-1 Drives LED Strings Up to 26V Compact 16-Lead TSSOP Thermally Enhanced Surface Mount Package
APPLICATIONS

Automotive and Avionic Lighting Architectural Detail Lighting Display Backlighting Constant Current Sources
L, LT, LTC and LTM are registered trademarks of Linear Technology Corporation. True Color PWM is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Patent Pending
TYPICAL APPLICATION
Step-Down 1A LED Driver
VIN 5V TO 36V 2.2F VIN SHDN LT3474 RT REF 80.6k 0.1F VADJ VC GND LED1 55 *SEE APPLICATIONS SECTION FOR DETAILS
3474 TA01a
Efficiency
95 0.22F EFFICIENCY (%) 10H 90 85 80 ONE WHITE 1A LED 75 70 65 60 0 200 400 800 600 LED CURRENT (mA) 1000
3474 G02
VIN = 12V
TWO SERIES CONNECTED WHITE 1A LEDS
BOOST SW BIAS OUT PWM LED
DIMMING* CONTROL
2.2F
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LT3474/LT3474-1 ABSOLUTE MAXIMUM RATINGS
(Note 1)
PIN CONFIGURATION
TOP VIEW DNC* OUT LED VIN SW BOOST BIAS GND 1 2 3 4 5 6 7 8 17 16 DNC* 15 GND 14 PWM 13 VADJ 12 VC 11 REF 10 SHDN 9 RT
VIN Pin ........................................................(-0.3V), 36V BIAS Pin....................................................................25V BOOST Pin Voltage ...................................................51V BOOST above SW Pin ...............................................25V OUT, LED Pins (LT3474) ............................................15V OUT, LED Pins (LT3474-1).........................................26V PWM Pin ...................................................................10V VADJ Pin .....................................................................6V , VC, REF RT Pins ..........................................................3V SHDN Pin ...................................................................VIN BIAS Pin Current .........................................................1A Maximum Junction Temperature (Note 2)............. 125C Operating Temperature Range (Note 3) LT3474E, LT3474E-1 ............................ -40C to 85C LT3474I, LT3474I-1 ............................ -40C to 125C Storage Temperature Range................... -65C to 150C Lead Temperature (Soldering, 10 sec) .................. 300C
FE PACKAGE 16-LEAD PLASTIC TSSOP JC = 8C/W, JA = 40C/W EXPOSED PAD (PIN 17) IS GND, MUST BE SOLDERED TO PCB *DO NOT CONNECT EXTERNAL CIRCUITRY TO THESE PINS.
ORDER INFORMATION
LEAD FREE FINISH LT3474EFE#PBF LT3474IFE#PBF LT3474EFE-1#PBF LT3474IFE-1#PBF TAPE AND REEL LT3474EFE#TRPBF LT3474IFE#TRPBF LT3474EFE-1#TRPBF LT3474IFE-1#TRPBF PART MARKING 3474EFE 3474IFE 3474EFE-1 3474IFE-1 PACKAGE DESCRIPTION 16-Lead TSSOP 16-Lead TSSOP 16-Lead TSSOP 16-Lead TSSOP TEMPERATURE RANGE -40C to 85C -40C to 125C -40C to 85C -40C to 125C
Consult LTC Marketing for parts specified with wider operating temperature ranges. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
ELECTRICAL CHARACTERISTICS
PARAMETER Minimum Input Voltage Input Quiescent Current Shutdown Current LED Pin Current
The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 12V, VBOOST = 16V, VOUT = 4V unless otherwise noted (Note 3).
CONDITIONS
l
MIN
TYP 3.5 2.6 0.01
MAX 4 4 2 1.02 1.025 0.207 0.210 1.265
UNITS V mA A A A A A V
Not Switching SHDN = 0.3V, VBOOST = 0V, VOUT = 0V VADJ Tied to VREF VADJ Tied to VREF/5
l l l
0.98 0.968 0.193 0.186 1.23
1 0.2 1.25
REF Voltage
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LT3474/LT3474-1 ELECTRICAL CHARACTERISTICS
PARAMETER Reference Voltage Line Regulation Reference Voltage Load Regulation VADJ Pin Bias Current (Note 4) Switching Frequency Maximum Duty Cycle RT = 80.6k RT = 80.6k RT = 10k RT = 232k RT = 80.6k, VOUT = 0V 2.6 VSHDN = SHDN Threshold 8.3 0.4 VC = 1V VC = 1V
The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 12V, VBOOST = 16V, VOUT = 4V unless otherwise noted (Note 3).
CONDITIONS 5V < VIN < 36V 0 < IREF < 250A
l l l
MIN
TYP 0.01 0.0002 20
MAX
UNITS %/V %/A
400 530 540
nA kHz kHz % % % kHz
470 450 90
500 95 76 98 70 2.65 10.3 0.9 0.8 100 100 1.5 1 2 1.9
Foldback Frequency SHDN Threshold (to Switch) SHDN Pin Current (Note 5) PWM Threshold VC Switching Threshold VC Source Current VC Sink Current LED to VC Current Gain LED to VC Transresistance VC to Switch Current Gain VC Clamp Voltage VC Pin Current in PWM Mode OUT Pin Clamp Voltage (LT3474) OUT Pin Current in PWM Mode Switch Current Limit (Note 6) Switch VCESAT Boost Pin Current Switch Leakage Current Minimum Boost Voltage (Note 7) Boost Diode Forward Voltage
2.7 12.3 1.2
V A V V A A A/mA V/mA A/V V
VC = 1V, VPWM = 0.3V VOUT = 4V, VPWM = 0.3V -40C to 85C LT3474I, LT3474I-1 at 125C ISW = 1A ISW = 1A
l
0.01 13.2 13.8 0.1 1.6 1.5 2.1 380 30 0.01 1.9
1 14.5 10 3.2 3.2 500 50 1 2.5
A V A A A mV mA A V mV
l l
IDIO = 100mA
600
Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. Note 3: The LT3474E and LT3474E-1 are guaranteed to meet performance specifications from 0C to 70C. Specifications over the -40C to 85C
operating temperature range are assured by design, characterization and correlation with statistical process controls. The LT3474I and LT3474I-1 are guaranteed to meet performance specifications over the -40C to 125C operating temperature range. Note 4: Current flows out of pin. Note 5: Current flows into pin. Note 6: Current limit is guaranteed by design and/or correlation to static test. Slope compensation reduces current limit at higher duty cycles. Note 7: This is the minimum voltage across the boost capacitor needed to guarantee full saturation of the internal power switch.
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LT3474/LT3474-1 TYPICAL PERFORMANCE CHARACTERISTICS
LED Current vs VADJ
1000 TA = 25C 800
1000 1200 VADJ = VREF
LED Current vs Temperature
700 600
Switch Voltage Drop
TA = 25C
SWITCH VOLTAGE DROP (mV)
100 125
LED CURRENT (mA)
LED CURRENT (mA)
800 600 400 200 0 -50 -25 VADJ = VREF/5
500 400 300 200 100 0 0
600
400
200
0
0
0.25
0.75 0.5 VADJ (V)
1
1.25
3474 GO3
50 25 75 0 TEMPERATURE (C)
1000 500 SWITCH CURRENT (mA)
1500
3474 G05
3474 G04
Current Limit vs Duty Cycle
2.5
Switch Current Limit vs Temperature
2.5
2.5
Current Limit vs Output Voltage
TA = 25C 2
2
CURRENT LIMIT (A)
2
MINIMUM (85C)
CURRENT LIMIT (A)
1.5
1.5
CURRENT LIMIT (A)
TYPICAL
1.5
MINIMUM (125C) 1
1
1
0.5
0.5
0.5
0
0
20
60 40 DUTY CYCLE (%)
80
100
3474 G06
0 -50
0
-25
50 25 0 75 TEMPERATURE (C)
100
125
0
2
4
6 VOUT (V)
8
10
12
3474 G08
3474 G07
Oscillator Frequency vs RT
600
TA = 25C
Oscillator Frequency vs Temperature
RT = 80.6k
Oscillator Frequency Foldback
600 TA = 25C RT = 80.6k
OSCILLATOR FREQUENCY (kHz)
OSCILLATOR FREQUENCY (kHz)
550
1000
OSCILLATOR FREQUENCY (kHz)
-25 75 0 25 50 TEMPERATURE (C) 100 125
500 400 300 200 100 0 0
500
450
100 10 RT (k)
3474 G09
100
400 -50
0.5
1
1.5 VOUT (V)
2
2.5
3474 G11
3474 G10
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LT3474/LT3474-1 TYPICAL PERFORMANCE CHARACTERISTICS
Boost Pin Current
60 TA = 25C 50
BOOST PIN CURRENT (mA) INPUT CURRENT (mA)
Quiescent Current
3.0 TA = 25C 2.5 2.0 1.5 1.0 0.5 0
Reference Voltage
1.260
1.255
40 30 20 10 0
VREF (V)
1.250
1.245
1.240
0
250
500 750 1000 1250 SWITCH CURRENT (mA)
1500
0
6
12
18 VIN (V)
24
30
36
3474 G13
1.235 -50
-25
50 25 0 75 TEMPERATURE (C)
100
125
3473 G12
3474 G14
Schottky Reverse Leakage
20
500
Schottky Forward Voltage Drop
TA = 25C
Open-Circuit Output Voltage and Input Current
60 TA = 25C 50
OUTPUT VOLTAGE (V)
VR = 5V
8 INPUT CURRENT LT3474-1 7
INPUT CURRENT (mA)
REVERSE CURRENT (A)
15
FORWARD CURRENT (mA)
400
6 5
40 LT3474 30 20 10 0 LT3474-1 LT3474 OUTPUT VOLTAGE
300
10
4 3 2 1 0 0 10 20 VIN (V) 30 40
3474 G16
200
5
100
0 -50
-25
75 0 25 50 TEMPERATURE (C)
100
125
0
0
200 600 800 400 FORWARD VOLTAGE (mV)
1000
3474 G19
3474 G15
Minimum Input Voltage, One White Luxeon III Star
6 5 4 TO RUN
VIN (V)
TA = 25C TO START
10
Minimum Input Voltage, Two Series Connected White Luxeon III Stars
TA = 25C
9
VIN (V)
LED VOLTAGE 3 2 1 0 0 200 400 600 800 LED CURRENT (mA) 1000
3474 G17
8
TO START
7 TO RUN LED VOLTAGE 6
5
0
200
600 800 400 LED CURRENT (mA)
1000
3474 G18
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LT3474/LT3474-1 PIN FUNCTIONS
DNC (Pins 1, 16): Do not connect external circuitry to these pins, or tie them to GND. Leave the DNC pins floating. OUT (Pin 2): The OUT pin is the input to the current sense resistor. Connect this pin to the inductor and the output capacitor. LED (Pin 3): The LED pin is the output of the current sense resistor. Connect the anode of the LED here. VIN (Pin 4): The VIN pin supplies current to the internal circuitry and to the internal power switch and must be locally bypassed. SW (Pin 5): The SW pin is the output of the internal power switch. Connect this pin to the inductor and switching diode. BOOST (Pin 6): The BOOST pin is used to provide a drive voltage, higher than the input voltage, to the internal bipolar NPN power switch. BIAS (Pin 7): The BIAS pin connects through a Schottky diode to BOOST. Tie to OUT. GND (Pins 8, 15, Exposed Pad Pin 17): Ground. Tie both GND pins and the Exposed Pad directly to the ground plane. The Exposed Pad metal of the package provides both electrical contact to ground and good thermal contact to the printed circuit board. It must be soldered to the circuit board for proper operation. RT (Pin 9): The RT pin is used to set the internal oscillator frequency. Tie an 80.6k resistor from RT to GND for a 500kHz switching frequency. SHDN (Pin 10): The SHDN pin is used to shut down the switching regulator and the internal bias circuits. The 2.6V switching threshold can function as an accurate under-voltage lockout. Pull below 0.3V to shut down the LT3474/LT3474-1. Pull above 2.65V to enable the LT3474/ LT3474-1. Tie to VIN if the SHDN function is unused. REF (Pin 11): The REF pin is the buffered output of the internal reference. Either tie the REF pin to the VADJ pin for a 1A output current, or use a resistor divider to generate a lower voltage at the VADJ pin. Leave this pin unconnected if unused. VC (Pin 12): The Vc pin is the output of the internal error amp. The voltage on this pin controls the peak switch current. Use this pin to compensate the control loop. VADJ (Pin 13): The VADJ pin is the input to the internal voltage to current amplifier. Connect the VADJ pin to the REF pin for a 1A output current. For lower output currents, program the VADJ pin using the following formula: ILED = 1A * VADJ/1.25V. PWM (Pin 14): The PWM pin controls the connection of the VC pin to the internal circuitry. When the PWM pin is low, the VC pin is disconnected from the internal circuitry and draws minimal current. If the PWM feature is unused, leave this pin unconnected.
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LT3474/LT3474-1 BLOCK DIAGRAM
VIN CIN BIAS SHDN INT REG AND UVLO 7 4 VIN
10
SLOPE COMP RT
R C1 S Q Q DRIVER
BOOST
6 C1
Q1 SW 5 D1
L1
9 RT
OSC
FREQUENCY FOLDBACK OUT 2 C2 100 0.1 LED 2V 3 DLED1
11
REF PWM
14
PWM VADJ Q2 USE WITH PWM DIMMING 12 CC1 CC2 RC 1.25k GND 8
3474 BD
13
+
1.25V gm VC
-
Figure 1. Block Diagram
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LT3474/LT3474-1 APPLICATIONS INFORMATION
Operation The LT3474 is a constant frequency, current mode regulator with an internal power switch capable of generating a constant 1A output. Operation can be best understood by referring to the Block Diagram. If the SHDN pin is tied to ground, the LT3474 is shut down and draws minimal current from the input source tied to VIN. If the SHDN pin exceeds 1.5V, the internal bias circuits turn on, including the internal regulator, reference, and oscillator. The switching regulator will only begin to operate when the SHDN pin exceeds 2.65V. The switcher is a current mode regulator. Instead of directly modulating the duty cycle of the power switch, the feedback loop controls the peak current in the switch during each cycle. Compared to voltage mode control, current mode control improves loop dynamics and provides cycle-bycycle current limit. A pulse from the oscillator sets the RS flip-flop and turns on the internal NPN bipolar power switch. Current in the switch and the external inductor begins to increase. When this current exceeds a level determined by the voltage at VC, current comparator C1 resets the flip-flop, turning off the switch. The current in the inductor flows through the external Schottky diode and begins to decrease. The cycle begins again at the next pulse from the oscillator. In this way, the voltage on the VC pin controls the current through the inductor to the output. The internal error amplifier regulates the output current by continually adjusting the VC pin voltage. The threshold for switching on the VC pin is 0.8V, and an active clamp of 1.9V limits the output current. The voltage on the VADJ pin sets the current through the LED pin. The NPN Q2 pulls a current proportional to the voltage on the VADJ pin through the 100 resistor. The gm amplifier servos the VC pin to set the current through the 0.1 resistor and the LED pin. When the voltage drop across the 0.1 resistor is equal to the voltage drop across the 100 resistor, the servo loop is balanced. Tying the REF pin to the VADJ pin sets the LED pin current to 1A. Tying a resistor divider to the REF pin allows the programming of LED pin currents of less than 1A. LED pin current can also be programmed by tying the VADJ pin directly to a voltage source up to 1.25V. An LED can be dimmed with pulse width modulation using the PWM pin and an external NFET. If the PWM pin is unconnected or pulled high, the part operates nominally. If the PWM pin is pulled low, the VC pin is disconnected from the internal circuitry and draws minimal current from the compensation capacitor. Circuitry drawing current from the OUT pin is also disabled. This way, the VC pin and the output capacitor store the state of the LED pin current until PWM is pulled high again. This leads to a highly linear relationship between pulse width and output light, allowing for a large and accurate dimming range. The RT pin allows programming of the switching frequency. For applications requiring the smallest external components possible, a fast switching frequency can be used. If very low or very high input voltages are required, a slower switching frequency can be programmed. During startup VOUT will be at a low voltage. The NPN Q2 can only operate correctly with sufficient voltage at VOUT, around 1.7V. A comparator senses VOUT and forces the VC pin high until VOUT rises above 2V, and Q2 is operating correctly. The switching regulator performs frequency foldback during overload conditions. An amplifier senses when VOUT is less than 2V and begins decreasing the oscillator frequency down from full frequency to 20% of the nominal frequency when VOUT = 0V. The OUT pin is less than 2V during startup, short circuit, and overload conditions. Frequency foldback helps limit switch current under these conditions.
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LT3474/LT3474-1 APPLICATIONS INFORMATION
The switch driver operates either from VIN or from the BOOST pin. An external capacitor and internal Schottky diode are used to generate a voltage at the BOOST pin that is higher than the input supply. This allows the driver to saturate the internal bipolar NPN power switch for efficient operation. Open Circuit Protection The LT3474 has internal open circuit protection. If the LED is absent or fails open, the LT3474 clamps the voltage on the LED pin at 14V. The switching regulator then skips cycles to limit the input current. The LT3474-1 has no internal open circuit protection. With the LT3474-1, be careful not to violate the ABSMAX voltage of the BOOST pin; if VIN > 25V, external open circuit protection circuitry (as shown in Figure 2) may be necessary. The output voltage during an open LED condition is shown in the Typical Performance Characteristics section. Undervoltage Lockout Undervoltage lockout (UVLO) is typically used in situations where the input supply is current limited, or has high source resistance. A switching regulator draws constant power from the source, so the source current increases as the source voltage drops. This looks like a negative resistance load to the source and can cause the source to current limit or latch low under low source voltage conditions. UVLO prevents the regulator from operating at source voltages where these problems might occur.
OUT 10k 27V VC
C1 R2 10.3A GND
An internal comparator will force the part into shutdown when VIN falls below 3.5V. If an adjustable UVLO threshold is required, the SHDN pin can be used. The threshold voltage of the SHDN pin comparator is 2.65V. A internal resistor pulls 10.3A to ground from the SHDN pin at the UVLO threshold. Choose resistors according to the following formula: R2 = 2.65V VTH - 2.65V - 10.3A R1
VTH = UVLO Threshold Example: Switching should not start until the input is above 8V. VTH = 8V R1 = 100k R2 = 2.65V = 61.9k 8V - 2.65V - 10.3A 100k
Keep the connections from the resistors to the SHDN pin short and make sure the coupling to the SW and BOOST pins is minimized. If high resistance values are used, the SHDN pin should be bypassed with a 1nF capacitor to prevent coupling problems from switching nodes.
LT3474 VIN R1 SHDN VIN 2.65V VC
100k
3474 F03
3474 F02
Figure 2. External Overvoltage Protection Circuitry for the LT3474-1.
Figure 3. Undervoltage Lockout
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LT3474/LT3474-1 APPLICATIONS INFORMATION
Setting the Switching Frequency The LT3474 uses a constant frequency architecture that can be programmed over a 200kHz to 2MHz range with a single external timing resistor from the RT pin to ground. The current that flows into the timing resistor is used to charge an internal oscillator capacitor. A graph for selecting the value of RT for a given operating frequency is shown in the Typical Performance Characteristics section. Table 1 shows suggested RT selections for a variety of switching frequencies.
Table 1. Switching Frequencies
SWITCHING FREQUENCY (MHz) 2 1.5 1 0.7 0.5 0.3 0.2 RT (k) 10 18.7 33.2 52.3 80.6 147 232
DC =
( VOUT + VF ) ( VIN - VSW + VF )
where VF is the forward voltage drop of the catch diode (~0.4V) and VSW is the voltage drop of the internal switch (~0.4V at maximum load). This leads to a minimum input voltage of: VIN(MIN) = VOUT + VF - VF + VSW DCMAX
with DCMAX = 1-tOFF(MIN) * f where t0FF(MIN) is equal to 200ns and f is the switching frequency. Example: f = 500kHz, VOUT = 4V DCMAX = 1- 200ns * 500kHz = 0.90 4V + 0.4V - 0.4V + 0.4V = 4.9 V VIN(MIN) = 9 0.9 The maximum operating voltage is determined by the absolute maximum ratings of the VIN and BOOST pins, and by the minimum duty cycle. VIN(MAX ) = VOUT + VF - VF + VSW DCMIN
Operating Frequency Selection The choice of operating frequency is determined by several factors. There is a tradeoff between efficiency and component size. Higher switching frequency allows the use of smaller inductors at the cost of increased switching losses and decreased efficiency. Another consideration is the maximum duty cycle. In certain applications, the converter needs to operate at a high duty cycle in order to work at the lowest input voltage possible. The LT3474 has a fixed oscillator off-time and a variable on-time. As a result, the maximum duty cycle increases as the switching frequency is decreased. Input Voltage Range The minimum operating voltage is determined either by the LT3474's undervoltage lockout of 4V, or by its maximum duty cycle. The duty cycle is the fraction of time that the internal switch is on and is determined by the input and output voltages:
with DCMIN = tON(MIN) * f where tON(MIN) is equal to 160ns and f is the switching frequency. Example: f = 500kHz, VOUT = 2.5V DCMIN = 160ns * 500kHz = 0.08 2.5V + 0.4V - 0.4V + 0.4V = 36 V VIN(MAX ) = 0 0.08 The minimum duty cycle depends on the switching frequency. Running at a lower switching frequency might allow a higher maximum operating voltage. Note that this is a restriction on the operating input voltage; the circuit will tolerate transient inputs up to the Absolute Maximum Rating.
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LT3474/LT3474-1 APPLICATIONS INFORMATION
Inductor Selection and Maximum Output Current A good first choice for the inductor value is 900kHz f where VF is the voltage drop of the catch diode (~0.4V), f is the switching frequency and L is in H. With this value the maximum load current will be 1.1A, independent of input voltage. The inductor's RMS current rating must be greater than the maximum load current and its saturation current should be at least 30% higher. For highest efficiency, the series resistance (DCR) should be less than 0.2. Table 2 lists several vendors and types that are suitable. For robust operation at full load and high input voltages (VIN > 30V), use an inductor with a saturation current higher than 2.5A. L = ( VOUT + VF ) *
Table 2. Inductors
PART NUMBER Sumida CR43-3R3 CR43-4R7 CDRH4D16-3R3 CDRH4D28-3R3 CDRH4D28-4R7 CDRH5D28-100 CDRH5D28-150 CDRH73-100 CDRH73-150 Coilcraft DO1606T-332 DO1606T-472 DO1608C-332 DO1608C-472 MOS6020-332 MOS6020-472 3.3 4.7 3.3 4.7 3.3 10 1.3 1.1 2 1.5 1.8 1.5 0.1 0.12 0.08 0.09 0.046 0.05 2 2 2.9 2.9 2 2 3.3 4.7 3.3 3.3 4.7 10 15 10 15 1.44 1.15 1.1 1.57 1.32 1.3 1.1 1.68 1.33 0.086 0.109 0.063 0.049 0.072 0.048 0.076 0.072 0.13 3.5 3.5 1.8 3 3 3 3 3.4 3.4 VALUE (H) IRMS (A) DCR () HEIGHT (mm)
The optimum inductor for a given application may differ from the one indicated by this simple design guide. A larger value inductor provides a higher maximum load current, and reduces the output voltage ripple. If your load is lower than the maximum load current, then you can relax the value of the inductor and operate with higher ripple current. This allows you to use a physically smaller inductor, or one with a lower DCR resulting in higher efficiency. Be aware that if the inductance differs from the simple rule above, then the maximum load current will depend on input voltage. In addition, low inductance may result in discontinuous mode operation, which further reduces maximum load current. For details of maximum output current and discontinuous mode operation, see Linear Technology's Application Note 44. Finally, for duty cycles greater than 50% (VOUT/VIN > 0.5), a minimum inductance is required to avoid sub-harmonic oscillations. See Application Note 19. The current in the inductor is a triangle wave with an average value equal to the load current. The peak switch current is equal to the output current plus half the peak-to-peak inductor ripple current. The LT3474 limits its switch current in order to protect itself and the system from overload faults. Therefore, the maximum output current that the LT3474 will deliver depends on the switch current limit, the inductor value, and the input and output voltages. When the switch is off, the potential across the inductor is the output voltage plus the catch diode drop. This gives the peak-to-peak ripple current in the inductor IL =
(1- DC) ( VOUT + VF ) (L * f)
where f is the switching frequency of the LT3474 and L is the value of the inductor. The peak inductor and switch current is ISW (PK ) =IL (PK ) =IOUT + IL 2
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LT3474/LT3474-1 APPLICATIONS INFORMATION
To maintain output regulation, this peak current must be less than the LT3474's switch current limit ILIM. For SW1, ILIM is at least 1.6A (1.5A at 125C) at low duty cycles and decreases linearly to 1.15A (1.08A at 125C) at DC = 0.8. The maximum output current is a function of the chosen inductor value: I IOUT (MAX ) = ILIM - L 2 =1.6A * (1 - 0.35 *DC) - IL 2 at the LT3474 input and to force this switching current into a tight local loop, minnimizing EMI. The input capacitor must have low impedance at the switching frequency to do this effectively, and it must have an adequate ripple current rating. The RMS input is: CINRMS = IOUT * VOUT VIN - VOUT VIN
(
) < IOUT
2
Choosing an inductor value so that the ripple current is small will allow a maximum output current near the switch current limit. One approach to choosing the inductor is to start with the simple rule given above, look at the available inductors, and choose one to meet cost or space goals. Then use these equations to check that the LT3474 will be able to deliver the required output current. Note again that these equations assume that the inductor current is continuous. Discontinuous operation occurs when IOUT is less than IL/2. Input Capacitor Selection Bypass the input of the LT3474 circuit with a 2.2F or higher ceramic capacitor of X7R or X5R type. A lower value or a less expensive Y5V type will work if there is additional bypassing provided by bulk electrolytic capacitors or if the input source impedance is low. The following paragraphs describe the input capacitor considerations in more detail. Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input capacitor is required to reduce the resulting voltage ripple
and is largest when VIN = 2VOUT (50% duty cycle). Considering that the maximum load current is 1A, RMS ripple current will always be less than 0.5A The high switching frequency of the LT3474 reduces the energy storage requirements of the input capacitor, so that the capacitance required is less than 10F The combination . of small size and low impedance (low equivalent series resistance or ESR) of ceramic capacitors makes them the preferred choice. The low ESR results in very low voltage ripple. Ceramic capacitors can handle larger magnitudes of ripple current than other capacitor types of the same value. Use X5R and X7R types. An alternative to a high value ceramic capacitor is a lower value ceramic along with a larger electrolytic capacitor. The electrolytic capacitor likely needs to be greater than 10F in order to meet the ESR and ripple current requirements. The input capacitor is likely to see high surge currents when the input source is applied. Tantalum capacitors can fail due to an over-surge of current. Only use tantalum capacitors with the appropriate surge current rating. The manufacturer may also recommend operation below the rated voltage of the capacitor.
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12
LT3474/LT3474-1 APPLICATIONS INFORMATION
A final caution is in order regarding the use of ceramic capacitors at the input. A ceramic input capacitor can combine with stray inductance to form a resonant tank circuit. If power is applied quickly (for example by plugging the circuit into a live power source), this tank can ring, doubling the input voltage and damaging the LT3474. The solution is to either clamp the input voltage or dampen the tank circuit by adding a lossy capacitor in parallel with the ceramic capacitor. For details, see Application Note 88. Output Capacitor Selection For most LEDs, a 2.2F 6.3V ceramic capacitor (X5R or X7R) at the output results in very low output voltage ripple and good transient response. Other types and values will also work; the following discusses tradeoffs in output ripple and transient performance. The output capacitor filters the inductor current to generate an output with low voltage ripple. It also stores energy in order to satisfy transient loads and stabilizes the LT3474's control loop. Because the LT3474 operates at a high frequency, minimal output capacitance is necessary. In addition, the control loop operates well with or without the presence of output capacitor series resistance (ESR). Ceramic capacitors, which achieve very low output ripple and small circuit size, are therefore an option. You can estimate output ripple with the following equation: VRIPPLE = IL (8 * f *COUT ) for ceramic capacitors
where IL is the peak-to-peak ripple current in the inductor. The RMS content of this ripple is very low so the RMS current rating of the output capacitor is usually not of concern. It can be estimated with the formula: IC (RMS) = IL 12
The low ESR and small size of ceramic capacitors make them the preferred type for LT3474 applications. Not all ceramic capacitors are the same, however. Many of the higher value capacitors use poor dielectrics with high temperature and voltage coefficients. In particular, Y5V and Z5U types lose a large fraction of their capacitance with applied voltage and at temperature extremes. Because loop stability and transient response depend on the value of COUT, this loss may be unacceptable. Use X7R and X5R types. Table 3 lists several capacitor vendors.
Table 3. Low-ESR Surface Mount Capacitors
VENDOR Taiyo-Yuden AVX TDK TYPE Ceramic Ceramic Ceramic SERIES X5R, X7R X5R, X7R X5R, X7R
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13
LT3474/LT3474-1 APPLICATIONS INFORMATION
Diode Selection The catch diode (D1 from Figure 1) conducts current only during switch off time. Average forward current in normal operation can be calculated from: ID( AVG) = IOUT ( VIN - VOUT ) VIN Table 4 lists several Schottky diodes and their manufacturers.
Table 4. Schottky Diodes
PART NUMBER On Semiconductor MBR0520L MBR0540 MBRM120E MBRM140 Diodes Inc. B0530W B120 B130 B140 HB International Rectifier 10BQ030 30 1 420 30 20 30 40 0.5 1 1 1 430 500 500 530 20 40 20 40 0.5 0.5 1 1 385 510 620 530 550 VR (V) I AVE (A) VF at 0.5A (mV) VF at 1A (mV)
The only reason to consider a diode with a larger current rating than necessary for nominal operation is for the worst-case condition of shorted output. The diode current will then increase to one half the typical peak switch current. Peak reverse voltage is equal to the regulator input voltage. Use a diode with a reverse voltage rating greater than the input voltage. If using the PWM mode of the LT3474, select a diode with low reverse leakage.
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14
LT3474/LT3474-1 APPLICATIONS INFORMATION
BOOST and BIAS Pin Considerations The capacitor and internal diode tied to the BOOST pin generate a voltage that is higher than the input voltage. In most cases, a 0.22F capacitor will work well. Figure 4 shows three ways to arrange the boost circuit. The BOOST pin must be more than 2.5V above the SW pin for full efficiency. For outputs of 2.8V or higher, the standard circuit (Figure 4a) is best. For lower output voltages, the BIAS pin can be tied to the input (Figure 4b). The circuit in Figure 4a is more efficient because the BOOST pin current comes from a lower voltage source. The BIAS pin can be tied to another source that is at least 3V (Figure 4c). For example, if a 3.3V source is on whenever the LED is on, the BIAS pin can be connected to the 3.3V output. For LT3474-1 applications with higher output voltages, an additional Zener diode may be necessary (Figure 4d) to maintain the BOOST pin voltage below the absolute maximum. In any case, be sure that the maximum voltage at the BOOST pin is both less than 51V and the voltage difference between the BOOST and SW pins is less than 25V. Programming LED Current
3474 F04a
VIN
BIAS BOOST LT3474 VIN SW GND VBOOST - VSW VOUT MAX VBOOST VIN + VOUT
C3 VOUT
(4a)
The LED current can be set by adjusting the voltage on the VADJ pin. For a 1A LED current, either tie VADJ to REF or to a 1.25V source. For lower output currents, program the VADJ using the following formula:
ILED = 1A * VADJ 1.25V
VIN
BIAS BOOST LT3474 VIN SW GND
C3 VOUT
3474 F04b
Voltages less than 1.25V can be generated with a voltage divider from the REF pin, as shown in Figure 5.
REF R1 VADJ LT3474 GND
3474 F04
VBOOST - VSW VIN MAX VBOOST 2VIN
(4b)
VIN2 > 3V BIAS BOOST LT3474 VIN SW GND
3474 F04c
C3 VOUT
R2
VIN
Figure 5. Setting VADJ with a Resistor Divider
VBOOST - VSW VIN2 MAX VBOOST VIN2 + VIN MINIMUM VALUE FOR VIN2 = 3V
(4c)
VIN
BIAS BOOST LT3474 VIN SW GND
C3 VOUT
In order to have accurate LED current, precision resistors are preferred (1% or better is recommended). Note that the VADJ pin sources a small amount of bias current, so use the following formula to choose resistors: R2 = VADJ 1.25V - VADJ + 50nA R1
3474 F04d
VBOOST - VSW VOUT - VZ MAX VBOOST VIN + VOUT - VZ
(4d)
Figure 4. Generating the Boost Voltage
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15
LT3474/LT3474-1 APPLICATIONS INFORMATION
To minimize the error from variations in VADJ pin current, use resistors with a parallel resistance of less than 4k. Use resistors with a series resistance of 5.11k or greater so as not to exceed the 250A current limit on the REF pin. Dimming Control There are several different types of dimming control circuits. One dimming control circuit (Figure 6) changes the voltage on the VADJ pin by tying a low on-resistance FET to the resistor divider string. This allows the selection of two different LED currents. For reliable operation, program an LED current of no less than 35mA. The maximum current dimming ratio (IRATIO) can be calculated from the maximum LED current (IMAX) and the minimum LED current (IMIN) as follows: IMAX = IRATIO IMIN Another dimming control circuit (Figure 7) uses the PWM pin and an external NFET tied to the cathode of the LED. When the PWM signal goes low, the NFET turns off, turning off the LED and leaving the output capacitor charged. The PWM pin is pulled low as well, which disconnects the VC pin, storing the voltage in the capacitor tied there. Use the C-RC string (tied to the VC pin) shown in Figure 7 for proper operation during start-up. When the PWM pin goes high again, the LED current returns rapidly to its previous on state since the compensation and output capacitors are at the correct voltage. This fast settling time allows The LT3474 to maintain diode current regulation with PWM pulse widths as short as 40s. If the NFET is omitted and the cathode of the LED is instead tied to GND, use PWM pulse widths of 1ms or greater. The maximmum PWM dimming ratio (PWMRATIO) can be calculated from the maximum PWM period (tMAX) and minimum PWM pulse width (tMIN) as follows: tMAX = PWMRATIO tMIN Total dimming ratio (DIMRATIO) is the product of the PWM dimming ratio and the current dimming ratio. Example: IMAX = 1A, IMIN = 0.1A, tMAX = 12ms, tMIN = 40s IRATIO = 1A =10:1 0.1A 12ms PWMRATIO = = 300:1 40s DIMRATIO =10 * 300 = 3000:1
REF R1 VADJ R2 DIM GND
3474 F05
LT3474
PWM 60Hz TO 10kHz
PWM LT3474 LED GND 3.3nF 10k 0.1F
3474 F06
Figure 6. Dimming with an NFET and Resistor Divider Figure 7. Dimming Using PWM Signal
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16
LT3474/LT3474-1 APPLICATIONS INFORMATION
LED Voltage Range The LT3474 can drive LED voltages from 2.4V to 12V. The LT3474-1 can drive LED voltages from 2.4V to 30V. Be careful not to exceed the ABSMAX rating of the OUT, LED, or BOOST pins of the LT3474-1 since the internal output clamp is disabled. See the Typical Application section for an example of adding an external output clamp. If the LED voltage can drift below 2.4V due to temperature or component variation, add extra series resistance to bring the overall voltage above 2.4V. Layout Hints As with all switching regulators, careful attention must be paid to the PCB layout and component placement. To maximize efficiency, switch rise and fall times are made as short as possible. To prevent electromagnetic interference (EMI) problems, proper layout of the high frequency switching path is essential. The voltage signal of the SW and BOOST pins have sharp rise and fall edges. Minimize the area of all traces connected to the BOOST and SW pins and always use a ground plane under the switching regulator to minimize interplane coupling. In addition, the ground connection for frequency setting resistor RT (refer to Figure 1) should be tied directly to the GND pin and not shared with any other component, ensuring a clean, noise-free connection.
PWM
SHDN
VIN
GND
VIA TO LOCAL GND PLANE VIA TO OUT
Figure 8. Recommended Component Placement
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17
LT3474/LT3474-1 TYPICAL APPLICATIONS
Step-Down 1A LED Driver with PWM Dimming
VIN 6V TO 36V C1 2.2F 50V VIN SHDN LT3474 RT REF R1 80.6k C4 3.3nF R2 10k C5 0.1F D1: B140HB C1 TO C3: X5R OR X7R M1: Si2302ADS VADJ VC GND LED1 M1 PWM BIAS OUT PWM LED C2 2.2F 6.3V
1ms/DIV ILED1 500mA/DIV
LED Current in PWM Mode
BOOST SW
C3 0.22F L1 10H 6.3V D1
V(PWM) 5V/DIV
3474 TA01
Step-Down 1A LED Driver with Two Series Connected LED Output
95
Efficiency, Two LED Output
VIN 12V TO 36V
90
SHDN LT3474 RT REF R1 33.2k C4 0.1F VADJ VC GND
SW D1 BIAS OUT PWM LED LED1 LED2 1A LED CURRENT C2 2.2F 10V
EFFICIENCY (%)
C1 2.2F 50V
VIN
BOOST
C3 0.22F L1 10H 10V
VIN = 12V VIN = 24V
85 80 75 70 65 60 55 0 200
400 800 600 LED CURRENT (mA)
1000
3474 G01
D1: MBRM 140 C1 TO C3: X5R OR X7R
3474 TA02
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18
LT3474/LT3474-1 PACKAGE DESCRIPTION
FE Package 16-Lead Plastic TSSOP (4.4mm)
(Reference LTC DWG # 05-08-1663)
Exposed Pad Variation BA
2.74 (.108)
4.90 - 5.10* (.193 - .201) 2.74 (.108) 16 1514 13 12 1110 9
6.60 0.10 4.50 0.10
SEE NOTE 4
2.74 (.108) 0.45 0.05 1.05 0.10 0.65 BSC 2.74 6.40 (.108) (.252) BSC
RECOMMENDED SOLDER PAD LAYOUT
12345678 1.10 (.0433) MAX
0 - 8
4.30 - 4.50* (.169 - .177)
0.25 REF
0.09 - 0.20 (.0035 - .0079)
0.50 - 0.75 (.020 - .030)
0.65 (.0256) BSC
NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS MILLIMETERS 2. DIMENSIONS ARE IN (INCHES) 3. DRAWING NOT TO SCALE
0.195 - 0.30 (.0077 - .0118) TYP
0.05 - 0.15 (.002 - .006)
FE16 (BA) TSSOP 0204
4. RECOMMENDED MINIMUM PCB METAL SIZE FOR EXPOSED PAD ATTACHMENT *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.150mm (.006") PER SIDE
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Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
19
LT3474/LT3474-1 TYPICAL APPLICATION
Step-Down 1A LED Driver with Four Series Connected LED Output
VIN 21V TO 36V C1 2.2F 50V C3 0.22F 16V D2 D1 L1 47H
VIN
BOOST
SHDN SW LT3474-1 BIAS RT REF OUT PWM LED GND
R1 80.6k C4 0.1F
VADJ VC
C2 2.2F 25V
R2 10k D3
12V TO 18V LED VOLTAGE 1A LED CURRENT fSW = 500kHz
Q1 R3 100k
3474 TA02a
D1: MBRM 140 D2: 7.5V Zener Diode D3: 22V Zener Diode Q1: MMBT3904 C1 TO C3: X5R OR X7R
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20 Linear Technology Corporation
(408) 432-1900 FAX: (408) 434-0507
LT 1008 REV D * PRINTED IN USA
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